Power controlling semiconductor device, switched-mode power supply, and method of designing the device and power-supply

ABSTRACT

A power controlling semiconductor device generates a drive pulse to turn on or off a switching device which applies an intermittent current to a primary side winding of a transformer in response to a voltage proportional to a present current flowing through the primary side winding of the transformer and to an output voltage detection signal from a secondary side of the transformer. The device includes a current detecting terminal and an overcurrent detecting circuit. The voltage proportional to the present current flowing through the primary side winding is applied to the current detecting terminal. The overcurrent detecting circuit compares a voltage corresponding to the voltage applied to the current detecting terminal with an upper limit current detecting voltage to detect an overcurrent. A correction current is applied to a correction resistor connected to the current detecting terminal for a shift in the voltage.

CROSS REFERENCE TO RELATED APPLICATION

This application is based upon and claims the benefit of priority under35 USC 119 of Japanese Patent Application No. 2018-084161 filed on Apr.25, 2018, the entire disclosure of which, including the description,claims, drawings and abstract, is incorporated herein by reference inits entirety.

BACKGROUND OF THE INVENTION 1. Field of the Invention

The present invention relates to a power controlling semiconductordevice, a DC power supply including such a device, and a method ofdesigning the device and the power supply. In particular, the presentinvention relates to a technique useful in correction of the operatingpoint of an overcurrent protection circuit and adjustment of outputripples in a primary circuit of a transformer disposed in an insulatedswitched-mode DC power supply.

2. Description of Related Art

A typical DC power supply is an insulated AC-DC converter consisting ofa diode bridge circuit that rectifies AC power into DC power and a DC-DCconverter that lowers the voltage of the DC power from the diode bridgecircuit to a desired potential. One of the known insulated AC-DCconverters includes a switched-mode power supply that turns on or off aswitching device connected in series to a primary side winding, forexample, by a pulse width modulation (PWM) controlling scheme or a pulsefrequency modulation (PFM) controlling scheme for control of the currentthrough the primary side winding and the induced voltage from asecondary side winding.

The switched-mode power supply has a predetermined rated load current ora maximum load current. Since an increase in current exceeding the ratedload current or an overcurrent in the secondary side may damage thepower supply in some cases, a primary control circuit may be providedthat has an overcurrent detecting function and overcurrent protectingfunction to turn off the switching device in response to the detectionof the overcurrent. JP 2005-341730A discloses a scheme for detection ofovercurrent at a load in a switched-mode power supply, where a resistorconnected in series to a primary switching device is disposed fordetection of the current and monitoring of the voltage converted fromcurrent at the resistor or the peak of the voltage having a triangularwaveform.

Traditional switched-mode power supplies usually have fixed rates ofoutput voltages. Meanwhile, power supplies capable of shifting theoutput voltages among, for example, 5V, 9V, 15V . . . , are desired inresponse to requirements on devices, such as USB power delivery (PD), atthe load.

For example, JP 2017-127109A discloses a switched-mode power supplyprovided with such a switching scheme.

In the case of the design of the switched-mode power supply that canswitch an output voltage, various ideas have been proposed on thedetermination of the output current corresponding to the output voltageduring the overcurrent protective operation (hereinafter referred to asan output overcurrent limit). For example, the output overcurrent limitmay be lowered as the output voltage increases or may be keptsubstantially constant despite a variation in output voltage.

Unfortunately, the switched-mode power supply disclosed in JP2017-127109A includes a controlling integrated circuit (IC) providedwith a switching device and an overcurrent protection controllingcircuit at a primary side. When the output voltage decreases, theovercurrent protection controlling circuit switches the overcurrentlimit of the switching device from a high reference voltage to a lowreference voltage and the maximum limit of the switching frequency froma high value to a low value. Thus, the output current during theovercurrent protective operation is determined in response to the outputvoltage.

In the case of the design of a switched-mode power supply including sucha controlling IC, a designer cannot freely determine the outputovercurrent limit after the determination of the output voltage. Theswitched-mode power supply in JP 2017-127109A inevitably determines alow switching frequency after a low output voltage is determined,resulting in high output ripples.

SUMMARY OF THE INVENTION

An object of the present invention, which has been made to solve theabove mentioned problem, is to provide a power controlling semiconductordevice of a switched-mode power supply that includes a transformerturning on or off the current flowing through a primary side winding foroutput control, where the relation between an output voltage and anoutput overcurrent limit can be freely varied when the output voltage isswitched in response to a request from a device at the load.

Another object of the present invention is to provide a powercontrolling semiconductor device that can switch the output voltagefreely determine the relation between the output voltage and an outputripple and to provide a switched-mode power supply including such adevice.

To achieve at least one of the abovementioned objects, according to anaspect of the present invention, a power controlling semiconductordevice generates a drive pulse to turn on or off a switching devicewhich applies an intermittent current to a primary side winding of atransformer in response to a voltage proportional to a present currentflowing through the primary side winding of the transformer and to anoutput voltage detection signal from a secondary side of thetransformer.

The device includes:

a current detecting terminal to which the voltage proportional to thepresent current flowing through the primary side winding is applied;

an overcurrent detecting circuit which compares a voltage correspondingto the voltage applied to the current detecting terminal with an upperlimit current detecting voltage to detect an overcurrent in thesecondary side of the transformer;

a turning-on signal generating circuit which generates a turning-onsignal for periodically turning on the switching device;

a turning-off signal generating circuit which generates a turning-offsignal for turning off the switching device in response to theovercurrent detected by the overcurrent detecting circuit; and

a correction current generating circuit which applies a correctioncurrent to the current detecting terminal, the correction currentcorresponding to the output voltage at the secondary side of thetransformer,

wherein the correction current generated by the correction currentgenerating circuit is applied to a correction resistor connected to thecurrent detecting terminal for a shift in the voltage applied to thecurrent detecting terminal.

The correction resistor may be an external component connected to thecurrent detecting terminal or a resistance switch disposed in asemiconductor chip.

In accordance with the power controlling semiconductor device having theconfiguration described above, the correction current is applied to thecorrection resistor and the voltage applied to the current detectingterminal is shifted thereby. The apparent upper limit current detectingvoltage can be thereby varied, and the correction current generated bythe correction current generating circuit shifts in response to theoutput voltage at the secondary side of the transformer. Hence, thevoltage applied to the current detecting terminal can vary in responseto the output voltage. Thus, the output overcurrent limit correspondingto the output voltage can be freely determined. As a result, multipleoutput voltages can be determined in, for example, a power supply inaccordance with the USB-PD standard. Appropriate output overcurrentlimits can be determined in response to the levels of the outputvoltages in the power supply.

Preferably, the power controlling semiconductor device further includes:

a control terminal which receives the output voltage detection signal;and

an overcurrent limiting voltage generator which generates apredetermined overcurrent limiting voltage,

wherein the turning-off signal generating circuit comprises:

-   -   a voltage selector which selects a lower one of a voltage        corresponding to the voltage at the control terminal and the        overcurrent limiting voltage as the upper limit current        detecting voltage; and    -   a voltage comparing circuit which compares the upper limit        current detecting voltage selected by the voltage selector with        the voltage corresponding to the voltage applied to the current        detecting terminal.

The “voltage corresponding to the voltage at the control terminal”includes, for example, a voltage at the control terminal, a voltageapplied through a buffer, a voltage divided or amplified from thevoltage at the control terminal, and a voltage applied through a slopecompensation circuit for prevention of subharmonic oscillation.

The “voltage corresponding to the voltage applied to the currentdetecting terminal” includes, for example, the voltage at the currentdetecting terminal, a voltage shifted at the current detecting terminalby the applied correction current through a correction resistor in asemiconductor chip, and a voltage amplified from the voltage at thecurrent detecting terminal or the voltage shifted from the voltage atthe current detecting terminal by a non-inverting amplifier.

In accordance with the configuration described above, the turn-offsignal for the switching device can be generated after the voltage atthe current detecting terminal reaches the voltage at the controlterminal in response to the voltage detection signal during the normaloperation. The turn-off signal for the switching device can be generatedafter the voltage at the current detecting terminal reaches theovercurrent limiting voltage during the overcurrent protectiveoperation. Thus, a seamless transition can be achieved between thenormal operation and the overcurrent protective operation.

Preferably, the transformer comprises a auxiliary winding, and thecorrection current generating circuit comprises:

a sample holding circuit which holds a voltage corresponding to avoltage induced in the auxiliary winding or a divided voltage from theinduced voltage during a demagnetization period of the transformer afterturning-off of the switching device; and

a voltage-current converting circuit which generates a currentcorresponding to the voltage held in the sample holding circuit.

The “voltage corresponding to” includes a voltage before a calculationby an arithmetic circuit, which will be described below in anembodiment, and a voltage after the calculation.

The configuration above can achieve an efficient correction currentgenerating circuit generating a desired correction current.

Preferably, the correction current generating circuit generates thecorrection current in response to the output voltage detection signal,the voltage induced in the auxiliary winding, or the divided voltagefrom the induced voltage, such that, if a level of the output voltagedetection signal increases from a first level to a second level, thecorrection current is adjusted to decrease the level of the outputvoltage detection signal to the first level or below.

In designing of a power supply having desired characteristics of theoutput current versus the switching frequency, the switching frequencyin a predetermined output current range can be reduced, resulting in anenhanced power conversion efficiency.

Preferably, the turning-on signal generating circuit comprises anoscillating circuit which generates an oscillation signal at apredetermined frequency, and

the oscillating circuit generates the oscillation signal at a frequencycorresponding to the voltage at the control terminal receiving theoutput voltage detection signal.

Free determination of the relation between the output voltage and theoutput ripple can achieve a power supply having a low output ripple. Asthe frequency increases at the predetermined output voltage and current,the ripple restriction effect is enhanced.

Preferably, a switched-mode power supply includes:

the power controlling semiconductor device;

a transformer;

a switching device which is connected in series to a primary sidewinding of the transformer;

a current-voltage converting resistor which is connected in series tothe switching device and applies a voltage converted from a current tothe current detecting terminal;

an output voltage detector which detects an output voltage at asecondary side of the transformer and transmits a detection signal tothe power controlling semiconductor device; and

a correction resistor which is connected between the current detectingterminal and one terminal of the current-voltage converting resistor.

A switched-mode power supply can be achieved that can vary the outputvoltage in response to a request from a device at the load andappropriately determine the output overcurrent limit.

According to another aspect of the present invention, a method ofdesigning a switched-mode power supply includes the power controllingsemiconductor device.

The method includes:

determining the number of windings and a voltage dividing ratio of afirst resistor to a second resistor such that an input voltage isapplied to a terminal and generates a first correction current at afirst output voltage and a second correction current at a second outputvoltage, an induced voltage from the auxiliary winding or a dividedvoltage which is divided from the induced voltage being applied to theterminal, the first resistor and the second resistor being connected tothe auxiliary winding;

determining a first current limit and a second current limit for theswitching device to generate a first output current at the first outputvoltage or a second output current at the second output voltage in thesecondary side of the transformer; and

determining resistances of a current-voltage converting resistor and acorrection resistor such that the resistances serve as the first andsecond current limits for the switching device if the first and secondoutput voltages and the upper limit current detecting voltage serve asovercurrent limiting voltages, the current-voltage converting resistorand the correction resistor being connected to the current detectingterminal.

Such a method can achieve the designing of a switched-mode power supplythat can vary the output voltage and has desired characteristics of theoutput overcurrent limit versus the output voltage.

According to still another aspect of the present invention, a method ofdesigning a switched-mode power supply includes the power controllingsemiconductor device.

The method includes:

determining a first upper limit current detecting voltage at a firstoutput voltage and a first output current at the secondary side of thetransformer;

determining the resistance of a current-voltage converting resistorconnected to the current detecting terminal such that the first upperlimit current detecting voltage is equal to the first output voltage atthe first output current in the secondary side of the transformer;

determining a second upper limit current detecting voltage equal to asecond output voltage at a second output current in the secondary sideof the transformer; and

determining the resistance of the correction resistor such that thesecond upper limit current detecting voltage is equal to the secondoutput voltage at the second output current in response to thecorrection current corresponding to the output voltage applied to thesecondary side of the transformer by a correction current generatingcircuit and such that the frequency for turning on or off a switchingdevice is held at or below a predetermined value.

Such a method can achieve the designing of a switched-mode power supplythat can vary the output voltage and has a low output ripple afterswitching to a low output voltage.

In accordance with the present invention, the power controllingsemiconductor device of a switched-mode power supply that includes atransformer turns on or off the current flowing through a primary sidewinding for output control, where the relation between the outputvoltage and the output overcurrent limit can be freely varied when theoutput voltage is switched in response to a request from a device at theload. The power controlling semiconductor device that can switch theoutput voltage freely determines the relation between the output voltageand the output ripple and to provide a switched-mode power supplyincluding such a device.

BRIEF DESCRIPTION OF THE DRAWINGS

The advantages and features provided by one or more embodiments of theinvention will become more fully understood from the detaileddescription given hereinbelow and the appended drawings which are givenby way of illustration only, and thus are not intended as a definitionof the limits of the present invention.

FIG. 1 illustrates a circuit configuration of a switched-mode powersupply or an AC-DC converter according to an embodiment of the presentinvention.

FIG. 2 is a block diagram illustrating the AC-DC converter in FIG. 1including a transformer and a switching power controlling circuit (powercontrolling IC) at a primary side according to a first embodiment.

FIG. 3A is a graph indicating a shift of a characteristic curve of anoutput overcurrent limit versus an output voltage during an overcurrentprotective operation after variations of the resistance of a correctionresistor of the AC-DC converter including the power controlling ICaccording to the first embodiment.

FIG. 3B is a graph indicating the shift of the characteristic curve ofthe output overcurrent limit versus the output voltage during theovercurrent protective operation after variations of resistances of thecorrection resistor and a current detecting resistor of the AC-DCconverter including the power controlling IC according to the firstembodiment.

FIG. 3C is a graph indicating the relation between the output voltageand a correction current in the power controlling IC according to thefirst embodiment.

FIG. 4 illustrates a specific circuit configuration of the powercontrolling IC according to the first embodiment.

FIG. 5 illustrates waveforms indicating a variation in voltage orcurrent at each component of the power controlling IC after the outputvoltage is adjusted to a high level in the first embodiment.

FIG. 6 illustrates waveforms indicating a variation in voltage orcurrent at each component of the power controlling IC after the outputvoltage is adjusted to a medium level in the first embodiment.

FIG. 7 illustrates waveforms indicating a variation in voltage orcurrent at each component of the power controlling IC after the outputvoltage is adjusted to a low level in the first embodiment.

FIG. 8 illustrate a circuit configuration of a variation in the powercontrolling IC according to the first embodiment.

FIG. 9A is a graph indicating characteristic curves of the correctioncurrent versus the control voltage at a correction current generatoraccording to an implementation of the first embodiment.

FIG. 9B is a graph indicating the characteristic curves of thecorrection current versus the control voltage at the correction currentgenerator according to the variation.

FIG. 9C is a graph indicating other exemplary characteristic curves ofthe correction current versus the control voltage at the correctioncurrent generator according to the variation.

FIG. 10 is a graph indicating characteristic curves of an upper limitcurrent detecting voltage versus the output voltage selected by avoltage selector according to the variation and characteristic curves ofa frequency Fsw of an oscillating circuit versus the control voltage.

FIG. 11A is a graph indicating characteristic curves of the switchingfrequency Fsw versus the output current according to the implementationof the first embodiment.

FIG. 11B is a graph indicating characteristic curves of the switchingfrequency Fsw versus the output current according to the variation.

FIG. 12 illustrates a circuit configuration of an exemplary correctioncurrent generator according to the variation in the power controlling ICof the first embodiment.

FIG. 13 is a block diagram of a power controlling IC according to asecond embodiment.

FIG. 14A is a graph indicating a shift of a characteristic curve of theswitching frequency versus the output current in the case that thecorrection resistor has a high resistance in the AC-DC converterincluding the power controlling IC according to the second embodiment.

FIG. 14B is a graph indicating a shift of the characteristic curve ofthe switching frequency versus the output current in the case that thecorrection resistor has a low resistance in the AC-DC converterincluding the power controlling IC according to the second embodiment.

FIG. 14C is a graph indicating the relation between the output voltageand the correction current in the power controlling IC according to thesecond embodiment.

FIG. 15 illustrates a specific circuit configuration of the powercontrolling IC according to the second embodiment.

FIG. 16 illustrates waveforms indicating a variation in voltage at eachcomponent of the power controlling IC after the output voltage isadjusted to a high level according to the second embodiment.

FIG. 17 illustrates waveforms indicating a variation in voltage at eachcomponent of the power controlling IC after the output voltage isadjusted to a low level according to the second embodiment.

FIG. 18A is a graph indicating a shift of a characteristic curve of thecontrol voltage versus the output current in the case of an correctionresistor has a high resistance in an AC-DC converter including a powercontrolling IC according to a third embodiment.

FIG. 18B is a graph indicating the shift of the characteristic curve ofthe control voltage versus the output current in the case of thecorrection resistor has a low resistance in the AC-DC converterincluding the power controlling IC according to the third embodiment.

FIG. 18C is a graph indicating the relation between the switchingfrequency and the control voltage in the power controlling IC accordingto the third embodiment.

FIG. 19 is a block diagram of the power controlling IC according to thethird embodiment.

FIG. 20 illustrates a specific circuit configuration of the powercontrolling IC according to the third embodiment.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

Preferred embodiments of the present invention will now be describedwith reference to the accompanying drawings.

First Embodiment

A switched-mode power supply according to a first embodiment can varyits output voltage in response to a request from a device at the loadand freely determine the relation between the output voltage and theoutput current during an overcurrent protective operation or an outputovercurrent limit.

FIG. 1 illustrates a circuit configuration of a switched-mode powersupply or an AC-DC converter according to the first embodiment of thepresent invention.

The AC-DC converter according to this embodiment includes a diode bridgecircuit 11 rectifying the AC voltage from a AC power source 10; acapacitor C1 smoothing the rectified voltage; a transformer 12 includinga primary side winding Np, a secondary side winding Ns, and a auxiliarywinding Nb, and a switching transistor SW including a N-type MOSFETconnected in series to the primary side winding Np of the transformer12; and a power controlling circuit 13 turning on or off the switchingtransistor SW.

In the present embodiment, the power controlling circuit 13 is asemiconductor integrated circuit in one semiconductor chip composed of,for example, a monocrystalline silicon (hereinafter referred to as“power controlling IC”) and includes an external terminal DRV outputtingsignals for turning on or off of the switching transistor SW to a gateterminal of the switching transistor SW.

At the secondary side of the transformer 12, a rectifying diode D2 isconnected in series to the secondary side winding Ns. A smoothingcapacitor C2 is connected between the cathode terminal of the diode D2and a ground point GND. The currents intermittently passing through theprimary side winding Np induces AC voltages in the secondary sidewinding Ns. The AC is rectified through the diode D2 and smoothed by thecapacitor C2, which then outputs a DC voltage Vout in proportional tothe turn ratio of the primary side winding Np to the secondary sidewinding Ns.

An output voltage detecting circuit 14 is also disposed at the secondaryside of the transformer 12 for detection of the output voltage Vout. Aphotocoupler (PC) 15 is disposed between the output voltage detectingcircuit 14 and the power controlling IC 13 to transmit an output voltagedetection signal in response to the voltage detected by the outputvoltage detecting circuit 14 to a control terminal CTRL of the powercontrolling IC 13.

The output voltage detecting circuit 14 includes resistors R21 and R22dividing the output voltage Vout and an error amplifier AMP receivingthe voltage divided by the resistors R21 and R22. The error amplifierAMP generates a voltage corresponding to the difference in potentialbetween a reference voltage at an output voltage switch 16 and thevoltage divided by the resistors R11 and R12. The output voltage switch16 can change the reference voltage to be generated in response to anoutput voltage switching instruction (a binary code or an analog signal)from a load device 17, for example, a USB device operating at the outputvoltage Vout of the AC-DC converter. The changed reference voltagevaries the output voltage of the error amplifier AMP and the outputvoltage detection signal feed backed to the primary side, resulting in avariation in the active time of the switching transistor SW by the powercontrolling IC 13. Thus, the output voltage Vout corresponding to thereference voltage is output from the AC-DC converter. The output voltageswitch may have any configuration that can vary the output voltage,besides that described above.

In the present embodiment, a current detecting resistor Rs is connectedbetween a source terminal of the switching transistor SW and a groundpoint GND. The current detecting resistor Rs applies a detection voltageVr to a current detecting terminal CS of the power controlling IC 13through a correction resistor Rcomp.

The AC-DC converter in this embodiment includes a rectifying/smoothingcircuit at the primary side. The rectifying/smoothing circuit includes arectifying diode DO connected in series to the auxiliary winding Nb anda smoothing capacitor C0 connected between the cathode terminal of thediode DO and the ground point GND. A voltage Vaux rectified and smoothedin the rectifying/smoothing circuit is applied to a power voltageterminal VDD of the power controlling IC 13. The terminals of theauxiliary winding Nb are connected to voltage dividing resistors R11 andR12, respectively. The resistors R11 and R12 divide a voltage Vvs andapply the divided voltage Vvs to a voltage detection terminal VS of thepower controlling IC 13.

An exemplary functional circuit configuration of the power controllingIC 13 according to the first embodiment will now be described withreference to FIG. 2.

As illustrated in FIG. 2, the power controlling IC 13 according to thepresent embodiment includes a turning-on signal generator 31 generatinga turning-on signal Son indicating the timing for turning on the primaryswitching transistor SW, for example, a clock signal having apredetermined frequency; and a turning-off signal generator 32 comparinga voltage Vcs at the current detecting terminal CS with an upper limitcurrent detecting voltage Vlim to generate a turning-off signal Soffindicating the timing for turning off the switching transistor SW.

The power controlling IC 13 includes a drive pulse generator 33generating a drive signal (drive pulse) Sdrv for the switchingtransistor SW in response to the turning-on signal Son generated by theturning-on signal generator 31 and the turning-off signal Soff generatedby the turning-off signal generator 32. The power controlling IC 13 alsoincludes a correction current generator 34 generating a correctioncurrent Icomp corresponding to a voltage Vvs at the voltage detectionterminal VS to apply the correction current Icomp to the currentdetecting terminal CS; and a current limit determiner 35 applying anupper limit current detecting voltage Vlim. The voltage Vvs at thevoltage detection terminal VS is generated by dividing the voltageinduced in the auxiliary winding Nb with the external series resistorsR11 and R12. The voltage induced in the auxiliary winding Nb isproportional to the sum of the output voltage Vout at the secondary sideof the AC-DC converter and a forward voltage VF at the diode D2. Theforward voltage VF is constant regardless of the output voltage Vout.Thus, the correction current Icomp corresponds to the output voltageVout.

The correction current Icomp generated by the correction currentgenerator 34 is applied from the current detecting terminal CS throughthe correction resistor Rcomp and the current detecting resistor Rs to aground point. The voltage at the current detecting terminal CS isthereby raised or shifted. The correction resistor Rcomp and the currentdetecting resistor Rs are external resistors. Thus, a user or designerof the IC can freely determine the resistances of the correctionresistor Rcomp and the current detecting resistor Rs and design aswitched-mode power supply with desired characteristics of the outputovercurrent limit versus the output voltage.

The characteristics of the output overcurrent limit versus the outputvoltage of the power controlling IC 13 after variations of theresistances of the correction resistor Rcomp and the current detectingresistor Rs and the overcurrent protective operation will now bedescribed.

As described above, the correction current Icomp generated by thecorrection current generator 34 passes through the correction resistorRcomp and raises the voltage at the current detecting terminal CS. Thus,the voltage Vcs increases as the correction current Icomp rises. Thus,the overcurrent protection in the power controlling IC with thecorrection resistor Rcomp and the overcurrent protection works at a lowoutput voltage Vout compared to a power controlling IC without thecorrection resistor Rcomp and the correction current generator 34. FIG.3A indicates the shift of a characteristic curve of the outputovercurrent limit versus the output voltage at a constant resistance ofthe current detecting resistor Rs and a varied resistance of thecorrection resistor Rcomp.

In FIG. 3A, the curve C0 indicates the characteristics of the outputovercurrent limit versus the output voltage where the correctionresistor Rcomp has a resistance of “zero”, in other words, acompensation voltage Vcomp is zero and thus no correction is performed.The curve C0 slopes down to the right. In other words, the outputovercurrent limit decreases as the output voltage Vout increases. Incontrast, the correction resistor Rcomp with a resistance of 2 kΩ or 4kΩ connected to the current detecting terminal CS produces acharacteristic curve C1 or C2 with a gentler slope, in FIG. 3A.

Variations in the resistances of the current detecting resistor Rs andthe correction resistor Rcomp can upwardly or downwardly shift thecharacteristic curve as illustrated by an arrow Y in FIG. 3A. As aresult, the variations in the resistances of the current detectingresistor Rs and the correction resistor Rcomp can achieve thecharacteristic curves illustrated in FIG. 3B.

FIG. 4 illustrates a specific circuit configuration of the powercontrolling IC 13 in FIG. 2.

As illustrated in FIG. 4, the turning-on signal generator 31, whichgenerates the turning-on signal Son in the power controlling IC 13 inthis embodiment, includes an oscillating circuit OSC, for example, aring oscillator oscillating at a predetermined frequency. The turning-onsignal generator 31 may include, for example, an oscillating circuitincluding a transducer and a divider dividing frequencies of oscillationsignals generated by the oscillating circuit. The turning-on signalgenerator 31 may be an oscillating circuit shifting the oscillatingfrequency in response to the control voltage Vctrl at the controlterminal CTRL.

The current limit determiner 35 includes an overcurrent limiting voltagegenerator 35 a, for example, a reference voltage circuit generating anovercurrent limiting voltage Vocp and a minimum selector 35 b selectingthe lower one of the generated overcurrent limiting voltage Vocp and thecontrol voltage Vctrl at the control terminal CTRL (or a voltage shiftedfrom the control voltage Vctrl and amplified by an amplifier). Theturning-off signal generator 32 generating the turning-off signal Soffincludes a voltage comparing circuit CMP comparing the upper limitcurrent detecting voltage Vlim (Vocp or Vctrl) with the voltage Vcs atthe current detecting terminal CS. Instead of the minimum selector 35 b,the current limit determiner 35 may include a first voltage comparingcircuit comparing the upper limit current detecting voltage Vlim withthe voltage Vcs at the current detecting terminal CS; a second voltagecomparing circuit comparing the control voltage Vctrl at the controlterminal CTRL with the voltage Vcs at the current detecting terminal CS;and a circuit selecting the earlier one of the signal from the firstvoltage comparing circuit and that from the second voltage comparingcircuit or an OR gate taking a logical add of the signals from the twovoltage comparing circuits.

In FIGS. 4, 8, and 15, the voltage Vcs at the current detecting terminalCS is directly applied to the voltage comparing circuit CMP.Alternatively, the applied voltage may preliminarily amplified by anamplifier (not shown) or shifted by a level shifter.

In FIGS. 4 and 8, the control voltage Vctrl at the control terminal CTRLis applied to the minimum selecting circuit 35 b. In FIG. 15, thecontrol voltage Vctrl is directly applied to the voltage comparingcircuit CMP. Alternatively, a voltage at a buffer (not shown), a voltageamplified or divided from a voltage at the control terminal, or avoltage of a slope compensating circuit for prevention of subharmonicoscillation may be applied.

The drive pulse generator 33 includes, for example, a RS flip-flop 33 ato be set by the turning-on signal Son generated by the turning-onsignal generator 31 and reset by the turning-off signal Soff generatedby the turning-off signal generator 32; and a drive circuit or driver 33b having a low output impedance that generates gate drive pulses Sdrvfor the switching transistor SW connected to the terminal DRV inresponse to an output from the flip-flop 33 a.

The correction current generator 34 includes, for example, a one-shotpulse generating circuit 34 a detecting a rise or fall in the outputfrom the flip-flop 33 a and generating a sampling signal Ssamp; a sampleholding circuit 34 b including a switch S1 to be turned on or off by thesampling signal Ssamp and a capacitor Ch to sample the voltage Vvs atthe voltage detection terminal VS; a voltage-current converting circuit34 c converting the sampled voltage into a current; and a switch S2 tobe turned on or off by the output from the flip-flop 33 a in a mannercomplementary to the switch S1 to apply the output current from thevoltage-current converting circuit 34 c to the current detectingterminal CS. The voltage-current converting circuit 34 c may include,for example, a transconductance (gm) amplifier receiving the sampledvoltage and a reference voltage Vref to generate a current correspondingto the difference in potential between the voltages.

As described above, the voltage Vvs at the voltage detection terminal VSreceiving the voltage divided from the induced voltage in the auxiliarywinding is proportional to the sum of the output voltage Vout at thesecondary side and the forward voltage VF at the diode D2. The outputvoltage Vout at the secondary side is shifted in response to a switchinginstruction from the load device. As illustrated in FIG. 3C, thevoltage-current converting circuit 34 c can generate a larger correctioncurrent Icomp as the output voltage Vout decreases or a smallercorrection current Icomp as the output voltage Vout increases where thegenerated correction current Icomp is applied to the current detectingterminal CS.

In the power controlling IC 13 according to the embodiment in FIG. 4,the overcurrent limiting voltage Vocp has a fixed value. Alternatively,the overcurrent limiting voltage Vocp may have a value associated withinput voltage Vin, a value associated with an ON-time of the switchingtransistor, or a value associated with the on-duty ratio of the drivepulse of the switching transistor SW. In this embodiment, the current inthe primary side winding is detected by the resistor Rs. Alternatively,the current may be detected by an on-resistance of the switchingtransistor SW, in other words, the voltage between the source and drainof the switching transistor SW.

In this embodiment, the turning-on signal Son indicating the timing forturning on the switching transistor SW is generated in the oscillatingcircuit OCS at a predetermined frequency. Alternatively, the turning-onsignal Son may be generated in response to the detection of the timewhen the current through the transformer 12 is zero; when the currentthrough the transformer 12 is zero and when the voltage between thesource and the drain of the switching transistor SW are zero; or whenthe resonant voltage at the switching transistor SW is dropped to anundetectable level.

The state of “the current through the transformer 12 is zero” may bedetected by the current through the secondary side winding, the voltageconverted from the current through the secondary side winding, theswitching transistor SW that is in OFF-state to completely demagnetizethe transformer 12 and generates an oscillating voltage, or theoscillation of the voltage at the secondary side winding or theauxiliary winding at the primary side. The turning-on signal Son may begenerated in response to such detection.

The overcurrent protecting operation of the power controlling IC 13 inFIG. 4 after a shift of the output voltage Vout will now be describedwith reference to timing charts in FIGS. 5 to 7. FIGS. 5, 6, and 7indicate operational timings at the output voltage Vout equaling Vout 1,Vout 2, and Vout 3, respectively, where Vout 1>Vout 2>Vout 3. Vfbrepresents a voltage sampled by the sample holding circuit 34 b. Idmgrepresents a shift in current through the secondary side winding of thetransformer 12 during a demagnetization period. The waveforms indicatinga shift of the voltage Vcs at the current detecting terminal CS in FIGS.5 to 7 demonstrate that the voltage Vcs reaching the voltage Vlim shiftsthe drive pulse Sdrv to a low level. Thus, the switching transistor SWis turned off.

As illustrated in FIGS. 5 to 7, the overcurrent protecting function ofthe power controlling IC 13 causes the voltage Vcs at the currentdetecting terminal CS to reach the upper limit current detecting voltageVlim (Vocp) and causes the gate drive pulse Sdrv to shift from a highlevel to a low level at times t1, t4, and t7, at which the switchingtransistor SW is turned off.

Thus, the voltage induced in the auxiliary winding rises, resulting in arise in the voltage Vvs at the voltage detection terminal VS. At timest1, t4, and t7, the sampling signal Ssamp shifts to a high level, whichturns on the switch S1 of the sample holding circuit 34 b and chargesthe capacitor Ch. The shift of the sampling signal Ssamp to a low levelat times t2, t5, and t8 causes the switch S1 to be turned off. Thevoltage Vvs at the terminal VS at any of the times is held in capacitorCh of the sample holding circuit 34 b as a voltage Vfb. Thevoltage-current converting circuit 34 c then generates a correctioncurrent Icomp inversely proportional to the held voltage Vfb. Thecorrection current Icomp is thereby applied to the current detectingterminal CS.

Since Vout 1>Vout 2>Vout 3, the correction current Icomp in FIG. 6 or 7is higher than that in FIG. 5. Thus, the voltage Vcomp raised by thecorrection current Icomp is also higher than that in FIG. 5. Asdemonstrated in FIGS. 6 and 7, the shift of the gate drive pulse Sdrv toa high level at times t3 and t6 causes the switching transistor SW to beturned on and the current to pass through the primary side winding.During a period Ton where the voltage Vcs at the current detectingterminal CS rises, the voltage Vcs reaches the upper limit currentdetecting voltage Vlim at a low current through the primary side windingcompared to the case where the correction current Icomp is not applied.During a normal operation at a low output current, the voltage Vcs atthe current detecting terminal CS reaching the control voltage Vctrl atthe control terminal CTRL instead of the overcurrent limiting voltageVocp causes the switching transistor SW to be turned off.

As a result, the current Idmg passing through the secondary side windingduring the demagnetization period decreases. As the low output voltageVout decreases, the apparent overcurrent limiting voltage for theovercurrent protection decreases. A variation in the resistance of thecorrection resistor Rcomp connected to the current detecting terminal CScan vary the voltage Vcomp raised by the correction current Icomp. Asillustrated in FIG. 3A, the slope of the characteristic curve of theoutput overcurrent limit in the overcurrent protective operation versusthe output voltage can be freely determined. The current detectingresistor Rs connected to the current detecting terminal CS has adifferent resistance from the correction resistor Rcomp. Thus, thecharacteristic curve of the output overcurrent limit versus the outputvoltage can be shifted as illustrated in FIG. 3B.

A specific scheme of determination of the characteristics of the outputovercurrent limit versus the output voltage during the overcurrentprotective operation of the power controlling IC 13 according to thepresent embodiment will now be described.

In the embodiment of FIG. 4, an output current Iout from the powercontrolling IC 13 of the switched-mode power supply in FIG. 1 can berepresented by Expression (i):

Iout=(Idmg1+Idmg2)×0.5×Tdmg/Tp  (i)

where Tp represents the switching cycle and Tdmg represents thedemagnetization period. The output current Iout is a time-averagedcurrents flowing through the secondary side winding of the transformerduring the demagnetization period.

The cycle Tp is the inverse of a frequency Fsw to be determined by theturning-on signal generator 31, i.e., Tp=1/Fsw. The relation between ademagnetized current Idmg1 and an ON-state current Ion2 at the switchingtransistor SW after turning-off of the switching transistor SW isrepresented by Expression (ii):

Idmg1=N1/N2×Ion2  (ii)

The relation is proportional and is determined by the ratio N1/N2 ofwindings of the primary side winding N1 to that of the secondary sidewinding N2.

The relation between a demagnetized current Idmg2 and the ON-statecurrent Ion 1 at the switching transistor SW after turning-on of theswitching transistor SW is represented by Expression (iii):

Idmg2=N1/N2×Ion1  (iii)

This relation is also proportional and is determined by the ratio N1/N2of windings of the primary side winding N1 to that of the secondary sidewinding N2.

The demagnetized current Idmg2 is represented by Expression (iv):

Idmg2=Idmg1−(Vout+VF)/L2×Tdmg  (iv)

The demagnetized current Idmg2 decreases from the demagnetized currentIdmg1 by the value determined by the output voltage Vout, the forwardvoltage VF at the diode D2, an inductance L2 at the secondary side ofthe transformer, and the demagnetization period Tdmg.

The ON-state current Ion2 is represented by Expression (v):

Ion2=Ion1+(Vin/L1)×Ton  (v)

The ON-state current Ion2 increases from the ON-state current Ion 1 bythe value determined by the input voltage Vin, an inductance L1 at theprimary side of the transformer, and an ON-time Ton.

The ON-state current Ion2 is represented by Expression (vi):

Ion2=(Vlim−Icomp×Rcomp)/Rs  (vi)

The ON-state current Ion2 is determined by the upper limit currentdetecting voltage Vlim, the correction current Icomp, the resistance ofthe correction resistor Rcomp, and the resistance of the currentdetecting resistor Rs.

The deformed expressions (i) to (vi) give Expression (vii):

$\begin{matrix}{{Iout} = {\left\{ {{{\left( {{Vlim} - {{Icomp} \times R\; 2}} \right)/R}\; 1 \times N\; {1/N}\; 2} - {{\left( {0.5 \times \left( {{Vout} + {VF}} \right)} \right)/\left( {L\; 2 \times {Fsw}} \right)} \times \left( {1 - {{\left( {{Vout} + {VF}} \right)/{Vin}} \times N\; {1/N}\; 2}} \right)}} \right\} \times \left( {1 - {{\left( {{Vout} + {VF}} \right)/{Vin}} \times N\; {1/N}\; 2}} \right)}} & ({vii})\end{matrix}$

Expression (vii) indicates the output current Iout as a function of theoutput voltage Vout, the correction current Icomp, the resistance of thecorrection resistor Rcomp, and the resistance of the current detectingresistor Rs. In the overcurrent state of the extremely high outputcurrent Iout, the overcurrent protecting function controls such that theupper limit current detecting voltage Vlim does not exceed theovercurrent limiting voltage Vocp. The output current Iout is limited bythe output overcurrent limit. Thus, a variation in the correctioncurrent Icomp in response to the voltage Vvs at the voltage detectionterminal VS or the output voltage Vout and the adjustment of theresistances of the correction resistor Rcomp and the current detectingresistor Rs can freely determine the characteristics of the outputovercurrent limit versus the output voltage during the overcurrentprotective operation.

In designing of a switched-mode power supply including a powercontrolling IC 13 having the configuration described above to determinedesired characteristics of an output overcurrent limit versus an outputvoltage, the number of windings of the auxiliary winding Nb and theresistances of the resistors R11 and R12 are determined such that thevoltage Vvs is applied to the voltage detection terminal VS. As aresult, the correction current at a first output voltage has a firstvalue and the correction current at a second output voltage has a secondvalue. The first and second current limits for the switching device arethen determined such that the output current in the secondary side ofthe transformer has a first output overcurrent limit at the first outputvoltage or a second output overcurrent limit at the second outputvoltage. In the case that the first and second output voltages and theupper limit current detecting voltage serves as overcurrent limitingvoltages, the resistance of a current-voltage converting resistorconnected to the current detecting terminal and the resistance of thecorrection resistor are determined such that the resistances serve asthe first and second current limits.

A variation in the power controlling IC 13 according to the firstembodiment will now be described with reference to FIGS. 8 to 12.

This variation differs from the embodiment described above in that theturning-on signal generator 31 includes an oscillating circuit OSCcapable of varying its frequency, in that the frequency varies inresponse to the control voltage Vctrl applied to the control terminalCTRL for generation of the turning-on signal Son, and in that thecorrection current generator 34 generates a correction current Icompcorresponding to the voltage Vvs at the voltage detection terminal VSand the control voltage Vctrl at the control terminal CTRL.

As illustrated by solid lines A in FIG. 10A, the oscillating circuit OSCcapable of varying the frequency generates the turning-on signal Son ata low frequency Fsw, for example, 20 kHz in an area of a low outputvoltage Vout where the control voltage Vctrl at the control terminalCTRL is, for example, 1.5 V or less. The oscillating circuit OSCgenerates the turning-on signal Son at a high frequency Fsw, for example65 kHz in an area of a high output voltage Vout where the controlvoltage Vctrl is, for example, higher than 1.7 V.

Switched-mode power supplies, as shown in the embodiment of FIG. 2, areoften designed to have an enhanced power conversion efficiency at aswitching frequency lower than that at a peak power output such that thepower conversion efficiency increases as the output power (the productof the output voltage and the output current) decreases relative to thepeak power output (the largest product of an output voltage and anoutput current selected from multiple combinations of rated outputvoltages and rated output currents). The reduced frequency Fsw of theoscillating circuit OSC in the area of the low output voltage Vout canthereby enhance the power conversion efficiency in that area compared tothe same frequency both in the area with the high output voltage Voutand in the area with the low output voltage Vout.

A variation in the frequency Fsw of the oscillating circuit OSC withoutthe correction current generator 34 in response to the control voltageVctrl at the control terminal CTRL produces a large shift in the outputcurrent during the overcurrent protective operation in response to theoutput voltage, resulting in a maximum difference of, for example, 3 Aor more. The oscillating circuit OSC with the correction currentgenerator 34 can raise the voltage Vcs at the current detecting terminalCS, resulting in a small shift in the output current during theovercurrent protective operation, for example a maximum difference of0.2 A or less.

In this variation, the control voltage Vctrl at the control terminalCTRL is applied to the minimum selecting circuit 35 b. As illustrated bydotted lines B in FIG. 10A, the minimum selecting circuit 35 bproportionally shifts the upper limit current detecting voltage Vlim tothe control voltage Vctrl at the control terminal CTRL, for example, bya factor of 0.208 in a normal operational area Ta where the outputcurrent Iout is lower than an output overcurrent limit Iocp. If thevoltage, which is proportional to the control voltage Vctrl, forexample, by a factor of 0.208, exceeds the overcurrent limiting voltageVocp and reaches, for example, 0.52 V, the minimum selecting circuit 35b can keep the upper limit current detecting voltage Vlim constant orequal to Vocp such that the overcurrent protecting function works, asillustrated in an operational area Tb.

The correction current generator 34 according to the embodiment in FIG.3 can generate a constant correction current Icomp regardless of thecontrol voltage Vctrl applied to the control terminal CTRL asillustrated in FIG. 9A. The correction current generator 34 according tothe present variation can generate a correction current Icompcorresponding to the control voltage Vctrl at the control terminal CTRLas illustrated in FIG. 9B.

In response to switching of the output voltage Vout to Vout 1, Vout 2,Vout 3, or Vout 4 (Vout 1>Vout 2>Vout 3>Vout 4), the correction currentgenerator 34 can generate the correction current Icomp according todifferent characteristic curves of the correction current correspondingto the output voltage versus the control voltage or the characteristicsof Icomp versus Vctrl. As illustrated in FIG. 9C, the correction currentgenerator 34 may be designed to vary the characteristic curves of thecorrection current versus the control voltage in a stepwise manner at apredetermined voltage (about 1.5 V in the drawing).

The power supply according to the embodiment in FIG. 2 varies the levelof the correction current Icomp according to the voltage Vvs at thevoltage detection terminal VS but keeps the control voltage Vctrl at thecontrol terminal CTRL constant (see FIG. 9A). As indicated by thecharacteristics of the switching frequency versus the output currentIout in FIG. 11A, the switching frequency is high at, for example, acurrent Iout of 1 A and Vout of 5 V compared to a current Iout of 1 Aand a voltage Vout of 20 V. The correction current generator 34 of thepower supply according to this variation alters the level of thecorrection current Icomp through the correction resistor Rcomp alsorelative to the control voltage Vctrl. As indicated by thecharacteristic curve of the switching frequency versus the outputcurrent Iout in FIG. 11B, the switching frequency is low at, forexample, a current Iout of 1 A and a voltage Vout of 5 V compared toIout of 1 A and Vout of 20 V. As a result, the power supply having adesired output overcurrent limit advantageously has a high degree offreedom in design and can achieve high power conversion efficiency at alow output voltage.

FIG. 12 illustrates a specific correction current generator 34 accordingto the present variation generating a correction current Icompcorresponding to the voltage Vvs at the voltage detection terminal VSand the control voltage Vctrl at the control terminal CTRL.

As illustrated in FIG. 12, the correction current generator 34 includesan arithmetic circuit 34 d receiving the voltage Vvs at the voltagedetection terminal VS and the control voltage Vctrl at the controlterminal CTRL. The voltage-current converting circuit 34 c can convertthe output voltage at the arithmetic circuit 34 d to a current andgenerate a correction current Icomp corresponding to the voltage Vvs atthe voltage detection terminal VS and the voltage Vctrl at the controlterminal CTRL. The sample holding circuit 34 b is disposed between thearithmetic circuit 34 d and the voltage-current converting circuit 34 cand includes two sets of sampling switches and capacitors to serve aslow-pass filters. It should be noted that the sample holding circuit 34b may be disposed at the preceding stage of the arithmetic circuit 34 d.

The arithmetic circuit 34 d includes a maximum selecting circuit SELselecting the higher one of the control voltage Vctrl at the controlterminal CTRL and a predetermined reference voltage Vref2; resistors R31and R32 dividing the output voltage at the maximum selecting circuitSEL; a voltage follower BUF converting the input having the voltage Vvsat the voltage detection terminal VS into an impedance-matched output;and a subtracting circuit SUB receiving the voltage divided from theoutput voltage at the maximum selecting circuit SEL and the outputvoltage at the voltage follower BUF. The voltage-current convertingcircuit 34 c generates a correction current Icomp corresponding to thedifference in potential between the voltage Vctrl at the controlterminal CTRL and the voltage Vvs at the voltage detection terminal VS.

Second Embodiment

A second embodiment of the switched-mode power supply according to thepresent invention will now be described.

FIG. 13 illustrates a circuit configuration of a switched-mode powersupply or a DC-DC converter according to the second embodiment.

The switched-mode power supply according to the second embodiment canswitch the output voltage in response to a request from a device at theload and freely determine the relation between the output voltage and anoutput ripple. Like the first embodiment in FIG. 2, the switched-modepower supply includes a correction current generator 34 generating acorrection current Icomp corresponding to the voltage at a voltagedetection terminal VS and applying the correction current Icomp to acurrent detecting terminal CS; and a correction resistor Rcomp connectedto a current detecting terminal CS.

The power controlling IC 13 in the second embodiment has two majordifferences from that in the first embodiment in FIG. 2.

The first difference is that the power controlling IC 13 in the firstembodiment applies the upper limit current detecting voltage Vlim to theturning-off signal generator 32 generating the turning-off signal Soffwhereas the power controlling IC 13 in the second embodiment applies thecontrol voltage Vctrl at the control terminal CTRL to the turning-offsignal generator 32.

The second difference is that the turning-on signal generator 31 of thepower controlling IC 13 in the first embodiment generates the turning-onsignal Son indicating the timing for turning on the switching transistorSW at a predetermined frequency whereas the turning-on signal generator31 of the power controlling IC 13 in the second embodiment generates theturning-on signal Son at the frequency corresponding to the controlvoltage Vctrl at the control terminal CTRL.

The operation of the power controlling IC 13 after variations inresistances of the correction resistor Rcomp and the current detectingresistor Rs according to the second embodiment will now be explained.

As described above, the output voltage shifted to a low level generatesa low switching frequency in a traditional switched-mode power supply asdisclosed in JP 2017-127109A, resulting in a high output ripple.

In the second embodiment, the correction current Icomp generated by thecorrection current generator 34 is applied to the correction resistorRcomp and raises the voltage at the current detecting terminal CS likethe first embodiment. As the correction current Icomp increases, thecurrent detecting voltage Vcs rises. The time for generation of theturning-off signal Soff can be advanced compared to the powercontrolling IC 13 without the correction resistor Rcomp and thecorrection current generator 34. Therefore, if the output voltage islow, the correction current Icomp is increased. This can prevent theswitching frequency from decreasing after selection of a low outputvoltage and prevent the output ripple from increasing.

FIG. 14A indicates a characteristic curve of the switching frequency andoutput current after connection of the correction resistor Rcomp ordetermination of a high resistance of the correction resistor Rcomp.FIG. 14B indicates a characteristic curve of the switching frequencyversus the output current without connection of the correction resistorRcomp or after determination of a low resistance of the correctionresistor Rcomp to vary the output voltage.

In each of FIGS. 14A and 14B, a solid line C1 represents thecharacteristic curve at the output voltage equal to Vout 1 and a dottedline C2 the characteristic curve at the output voltage equal to Vout 2,where Vout 1>Vout 2. In the absence of the correction resistor Rcomp orat a low resistance of the correction resistor Rcomp, the increasingoutput voltage Vout 2 cannot readily increase the switching frequencyFsw despite an increase in the output current Iout, as indicated by thecharacteristic curve C2 in FIG. 14B.

In contrast, as the output current Iout increases both at the high andlow output voltages as indicated by the characteristic curve C2 in FIG.14A, a high resistance of the correction resistor Rcomp increases theswitching frequency Fsw. As a result, selection of a low output voltagedoes not generate a high output ripple. Furthermore, a variation in theresistance of the external correction resistor Rcomp can shift thecharacteristic curve C2 in FIG. 14A.

In this manner, the switched-mode power supply according to the secondembodiment can freely determine the relation between the output voltageand the output ripple. The output ripple can be also adjusted by theswitching frequency or the frequency of the oscillating circuit. As thefrequency increases, the output ripple can decrease. At the same time, ahigh frequency is likely to lower the power conversion efficiency ingeneral. Thus, the frequency of the oscillating circuit should bedetermined in view of the trade-off relation between the output rippleand the power conversion efficiency.

FIG. 15 illustrates a specific circuit configuration of the powercontrolling IC 13 in FIG. 13.

As illustrated in FIG. 15, the power controlling IC 13 according to thisembodiment includes a turning-on signal generator 31 generating theturning-on signal Son and including an oscillating circuit OSC thatvaries the frequency according to the control voltage Vctrl at thecontrol terminal CTRL. The oscillating circuit OSC may operate at aconstant frequency.

The turning-off signal generator 32 generating the turning-off signalSoff includes a voltage comparing circuit CMP comparing the controlvoltage Vctrl at the control terminal CTRL with the voltage Vcs at thecurrent detecting terminal CS. Like the first embodiment in FIG. 4, theturning-off signal generator 32 may include an overcurrent limitingvoltage generator 35 a generating the overcurrent limiting voltage Vocp;a minimum selecting circuit 35 b selecting the lower one of theovercurrent limiting voltage Vocp and the control voltage Vctrl at thecontrol terminal CTRL as the upper limit current detecting voltage Vlim;and a voltage comparing circuit CMP comparing the upper limit currentdetecting voltage Vlim (Vocp or Vctrl) with the voltage Vcs at thecurrent detecting terminal CS.

The drive pulse generator 33 and the correction current generator 34have the same configurations as those in the first embodiment in FIG. 4and are not described.

Also in this embodiment, the output voltage Vout at the secondary sidecan vary in response to a switching instruction from a load device. Asillustrated in FIG. 14C, the voltage-current converting circuit 34 c ofthe correction current generator 34 can generate an larger correctioncurrent Icomp as the output voltage Vout decreases or a smallercorrection current Icomp as the output voltage Vout increases where thegenerated correction current Icomp is applied to the current detectingterminal CS.

The power controlling IC 13 according to the embodiment in FIG. 15generates the turning-on signal Son indicating the timing for turning onthe switching transistor SW in the oscillating circuit OSC at afrequency variable according to the control voltage Vctrl at the controlterminal CTRL. Alternatively the power controlling IC 13 may generate aturning-on signal Son in response to the detection of the time when thecurrent through the transformer 12 is zero; the time when the currentthrough the transformer 12 is zero and when the voltage between thesource and the drain of the switching transistor SW is zero; or when theoscillating voltage at the switching transistor SW is dropped to anundetectable level, with the proviso that a time corresponding to thecontrol voltage Vctrl at the control terminal CTRL elapses.

The state of “the current through the transformer 12 is zero” may bedetected by the current through the secondary side winding, the voltageconverted from the current through the secondary side winding, theswitching transistor SW that is in OFF-state to completely demagnetizethe transformer 12 and generates an oscillating voltage, or theoscillation of the voltage at the secondary side winding or theauxiliary winding at the primary side. The turning-on signal Son may begenerated in response to such detection.

The operation of the power controlling IC 13 in FIG. 15 after a shift ofthe output voltage Vout will now be explained with reference to FIGS. 16and 17. FIGS. 16 and 17 indicate the operational timings of theswitching transistor SW at the output voltage Vout equal to Vout 1 andVout 2, respectively, where Vout 1>Vout 2; and the output current Ioutequal to Iout 1. Vfb represents a voltage sampled by the sample holdingcircuit 34 b. Idmg represents a shift in current through the secondaryside winding of the transformer 12 during a demagnetization period. Thewaveforms indicating a shift of the voltage Vcs at the current detectingterminal CS demonstrate that the switching transistor SW is turned offafter the voltage Vcs reaches the control voltage Vctrl at the controlterminal CTRL.

As illustrated in FIGS. 16 and 17, the power controlling IC 13 causesthe voltage Vcs at the current detecting terminal CS at the secondaryside to reach the control voltage Vctrl applied to the control terminalCTRL and causes the gate drive pulse Sdrv to shift from a high level toa low level at times t1, t4, and t7, at which the switching transistorSW is turned off. The voltage induced in the auxiliary winding therebyrises, resulting in a rise in a voltage Vvs at the voltage detectionterminal VS. At times t1, t4, and t7, a sampling signal Ssamp shifts toa high level, which turns on a switch S1 of the sample holding circuit34 b and charges the capacitor Ch. The shift of the sampling signalSsamp to a low level at the times t2, t5, and t8 causes the switch S1 tobe turned off. The voltage Vvs at the terminal VS is held in capacitorCh of the sample holding circuit 34 b as a voltage Vfb. Thevoltage-current converting circuit 34 c then generates a correctioncurrent Icomp inversely proportional to the held voltage Vfb. Thecorrection current Icomp is thereby applied to the current detectingterminal CS.

Since Vout 1>Vout 2, the correction current Icomp in FIG. 17 is higherthan that in FIG. 16. Thus, the voltage Vcomp raised by the correctioncurrent Icomp in FIG. 17 is also higher than that in FIG. 16. The shiftof the gate drive pulse Sdrv to a high level at times t3 and t6 causesthe switching transistor SW to be turned on and a the current to passthrough the primary side winding. During a period Ton where the voltageVcs at the current detecting terminal CS rises, the voltage Vcs reachesthe control voltage Vctrl at the secondary side at a low detectionvoltage Vr or a peak current Ion2 compared to a low correction currentIcomp. The output voltage Vout is thereby lower than Vout 2. The outputvoltage detector 14 controls the voltage Vctrl such that Vout=Vout 2,resulting in increases in the control voltage Vctrl and the switchingfrequency. In other words, the switching frequency decreases.

If the correction current Icomp or the resistance of the correctionresistor Rcomp were zero, the switching frequency would increase,resulting in a high output ripple.

However, the switching frequency decreases in this embodiment asdescribed above, resulting in a low output ripple.

A scheme of determination of the characteristics of the switchingfrequency versus the output current in the power controlling IC 13according to the present embodiment will now be described.

An output current Iout from the switched-mode power supply in FIG. 10including the power controlling IC 13 according to the embodiment inFIG. 10 can be represented by Expression (viii):

Iout=(Idmg1+Idmg2)×0.5×Tdmg/Tp  (viii)

where Tp represents the switching cycle and Tdmg represents thedemagnetization period. The output current Iout is a time-averagedcurrent flowing through the secondary side winding of the transformerduring a demagnetization period.

The cycle Tp is the inverse of a frequency Fsw to be determined by theovercurrent protecting function, i.e., Tp=1/Fsw. The demagnetizationperiod Tdmg is represented by Expression (ix):

Tdmg=(L2×Idmg1)/(Vout+VF)  (ix)

The demagnetization period Tdmg is proportional to a peak current Idmg1in the secondary side winding of the transformer and an inductance L2 inthe secondary side winding of the transformer and is inverselyproportional to the sum of the output voltage Vout and the forwardvoltage VF at the diode D2.

The relation between the demagnetized current Idmg1 and the peak currentIon2 through the switching transistor SW after turning-off of theswitching transistor SW is represented by Expression (x):

Idmg1=(N1/N2)×Ion2  (x)

The relation is proportional and is determined by the ratio N1/N2 ofwindings of the primary side winding N1 to windings of the secondaryside winding N2.

The peak current Ion2 through the switching transistor SW is representedby Expression (xi):

Ion2=(Vctrl−Rcomp×Icomp)/Rs  (xi)

The peak current Ion2 is determined by the control voltage Vctrl at thecontrol terminal CTRL, the resistance of the current detecting resistorRs, the correction current Icomp, and the resistance of the correctionresistor Rcomp.

The inductance ratio of the primary side winding to the secondary sidewinding L1/L2 is represented by Expression (xii):

L1/L2=(N1/N2)²  (xii)

The inductance ratio is the square of the ratio N1/N2 of windings.

To consolidate the expressions (viii) to (xii), the switching frequencyFsw is represented by Expression (xiii):

$\begin{matrix}{{Fsw} = {{\left( {2 \times \left( {{Vout} + {VF}} \right) \times {Iout}} \right)/L}\; 1 \times \left( {{Rs}/\left( {{Vctrl} - {{Rcomp} \times {Icomp}}} \right)} \right)^{2}}} & ({xiii})\end{matrix}$

The switching frequency Fsw is correlated with the output voltage Vout,the output current Iout, the control voltage Vctrl at the controlterminal CTRL, the resistance of the current detecting resistor Rs, andthe correction current Icomp.

Supposing that the oscillating circuit OSC is designed so as todetermine the relation between the control voltage Vctrl and thefrequency Fsw of the oscillating circuit OSC as illustrated in FIG. 18C,the current detecting resistor Rs has, for example, a constantresistance whereas the correction resistor Rcomp has a high resistance,for example, 4 kQ. Thus, the relation between the control voltage Vctrland the output current Iout can be determined as illustrated in FIG.18A, resulting in the characteristic curves of the switching frequencyversus the output current in FIG. 14A. A design that establishes therelation between the control voltage Vctrl and the output current Ioutin FIG. 18B can achieve the characteristic curves of the switchingfrequency versus the output current in FIG. 14B.

In each of FIGS. 18A and 18B, the solid line C1 represents acharacteristic curve of the output voltage equal to Vout 1 and thedotted line C2 a characteristic curve of the output voltage equal toVout 2, where Vout 1>Vout 2.

A switched-mode power supply that includes the power controlling IC 13having the configuration described above and that has desiredcharacteristics and a low ripple is designed as follows. A first upperlimit current detecting voltage at a first output voltage and a firstoutput current at the secondary side of the transformer is determined.If the output voltage is a first output voltage and the output currentis a first output current in the secondary side of the transformer, theresistance of a current-voltage converting resistor connected to thecurrent detecting terminal is determined such that the first upper limitcurrent detecting voltage is equal to the first output voltage. If theoutput voltage is a second output voltage and the output current is asecond output current in the secondary side of the transformer, a secondupper limit current detecting voltage is determined. The resistance ofthe correction resistor is determined such that the second upper limitcurrent detecting voltage is equal to the second output voltage at thesecond output current in response to the correction current from acorrection current generating circuit corresponding to the outputvoltage applied to the secondary side of the transformer and such thatthe frequency for turning on or off a switching device is held at orbelow a predetermined value. A ripple restriction effect can be acquiredin the case that the frequency of the switching device does not vary orincreases as the intensity of the output voltage detection signalincreases.

Third Embodiment

A third embodiment of a switched-mode power supply according to thepresent invention will now be described.

FIG. 19 illustrates a circuit configuration of a switched-mode powersupply or a DC-DC converter according to the third embodiment of thepresent invention.

The switched-mode power supply according to the third embodiment is acombination of those of the first and second embodiments. The powersupply can vary the output voltage in response to a request from adevice at the load side and freely determine the relation between theoutput voltage and the output current during the overcurrent protectiveoperation and the relation between the output voltage and the outputripple. As in the first embodiment in FIG. 2, the switched-mode powersupply includes a correction current generator 34 generating acorrection current Icomp corresponding to the voltage at the voltagedetection terminal VS and applying the correction current Icomp to thecurrent detecting terminal CS; and a correction resistor Rcomp connectedto the current detecting terminal CS.

The switched-mode power supply in the third embodiment has two majordifferences from that in the first embodiment in FIG. 2, as follows:

First, the power controlling IC of the power supply in the firstembodiment applies the upper limit current detecting voltage Vlim fromthe current limit determiner 35 to the turning-off signal generator 32generating the turning-off signal Soff whereas the power controlling ICof the power supply in the third embodiment selects one of theovercurrent limiting voltage Vocp and the control voltage Vctrl appliedto the control terminal CTRL as the upper limit current detectingvoltage Vlim and applies the selected voltage to the current limitdeterminer 35.

Second, the turning-on signal generator 31 of the power controlling ICin the first embodiment generates the turning-on signal Son indicatingthe timing for turning on the switching transistor SW at a predeterminedfrequency whereas the turning-on signal generator 31 of the powercontrolling IC in the third embodiment generates the turning-on signalSon at a frequency corresponding to the control voltage Vctrl at thecontrol terminal CTRL.

FIG. 20 illustrates a specific circuit configuration of the powercontrolling IC in FIG. 19.

As illustrated in FIG. 20, the turning-on signal generator 31, whichgenerates the turning-on signal Son in the power controlling IC 13according to this embodiment, includes an oscillating circuit OSC thefrequency of which varies in response to the control voltage Vctrl atthe control terminal CTRL.

The current limit determiner 35 includes an overcurrent limiting voltagegenerator 35 a generating an overcurrent limiting voltage Vocp; and aminimum selecting circuit 35 b selecting the lower one of theovercurrent limiting voltage Vocp and the control voltage Vctrl at thecontrol terminal CTRL. The turning-off signal generator 32 generatingthe turning-off signal Soff includes a voltage comparing circuit CMPcomparing the voltage Vlim selected by the minimum selecting circuit 35b, in other words, the overcurrent limiting voltage Vocp or the controlvoltage Vctrl with the voltage Vcs at the current detecting terminal CS.

The drive pulse generator 33 includes, for example, a RS flip-flop 33 ato be set by the turning-on signal Son generated from the turning-onsignal generator 31 and reset by the turning-off signal Soff generatedfrom the turning-off signal generator 32; and a drive circuit or driver33 b generating a gate drive pulse Sdrv for the switching transistor SWconnected to a terminal DRV in response to the output from the flip-flop33 a.

The correction current generator 34 includes, for example, a one-shotpulse generating circuit 34 a detecting a rise or fall in the outputfrom the flip-flop 33 a and generating a sampling signal Ssamp; a sampleholding circuit 34 b including a switch S1 to be turned on or off by thesampling signal Ssamp and a capacitor Ch to sample the voltage Vvs atthe voltage detection terminal VS; a voltage-current converting circuit34 c converting the sampled voltage into a current; and a switch S2 tobe turned on or off by the output from the flip-flop 33 a in a mannercomplementary to the switch S1 to apply the output current from thevoltage-current converting circuit 34 c to the current detectingterminal CS.

In response to a switching instruction from a load device, the outputfrom the output voltage detecting circuit 14 (see FIG. 1) varies. Thecontrol voltage Vctrl at the control terminal CTRL is varied thereby,which can vary the output voltage Vout at the secondary side. Thevoltage-current converting circuit 34 c increases the correction currentIcomp as the output voltage Vout decreases or decreases the correctioncurrent Icomp as the output voltage Vout increases. The correctioncurrent Icomp is applied to the current detecting terminal CS.

In this embodiment, the resistance of the correction resistor Rcompconnected to the current detecting terminal CS or the resistances of thecorrection resistor Rcomp and the current detecting resistor Rs areappropriately determined, resulting in desired characteristic curves ofthe output current versus the output voltage when the overcurrentprotecting function works as indicated in FIGS. 3A and 3B and desiredcharacteristic curves of the switching frequency versus the outputcurrent as indicated in FIGS. 18A and 18B. The adjustment of onecharacteristic by a variation in resistance of the correction resistorRcomp varies the other characteristics. The characteristics should beprioritized and be balanced.

Although the invention made by the inventor has been specificallydescribed with reference to the embodiments, the invention is notlimited to the embodiments. For example, the switching transistor SWapplying an intermittent current to the primary side winding of thetransformer in the present embodiments is a separate component from thepower controlling IC 13. Alternatively, the switching transistor SW maybe integrated with the power controlling IC 13 into one semiconductorintegrated circuit.

In the present embodiments, the correction resistor Rcomp and thecurrent detecting resistor Rs are disposed outside the power controllingIC 13. Alternatively, the correction resistor Rcomp and/or the currentdetecting resistor Rs may be disposed as a variable resistor(s) in thepower controlling IC 13, and the variable resistance (s) may beappropriately determined by a laser or a mask option.

In the embodiments, the diode D2 is connected in series to the secondaryside winding Ns to serve as a rectifier rectifying the AC voltageinduced in the secondary side winding Ns. Alternatively, a differentrectifying circuit, for example, a synchronous rectifier circuit may beconnected in series to the secondary side winding Ns.

In the embodiments, the AC-DC converter including a power controlling ICaccording to the present invention is of a flayback type. Alternatively,the AC-DC converter including the power controlling IC according to thepresent invention may be of a forward type or a quasi-resonant type.

What is claimed is:
 1. A power controlling semiconductor device forgeneration of a drive pulse to turn on or off a switching device whichapplies an intermittent current to a primary side winding of atransformer in response to a voltage proportional to a present currentflowing through the primary side winding of the transformer and to anoutput voltage detection signal from a secondary side of thetransformer, the device comprising: a current detecting terminal towhich the voltage proportional to the present current flowing throughthe primary side winding is applied; an overcurrent detecting circuitwhich compares a voltage corresponding to the voltage applied to thecurrent detecting terminal with an upper limit current detecting voltageto detect an overcurrent in the secondary side of the transformer; aturning-on signal generating circuit which generates a turning-on signalfor periodically turning on the switching device; a turning-off signalgenerating circuit which generates a turning-off signal for turning offthe switching device in response to the overcurrent detected by theovercurrent detecting circuit; and a correction current generatingcircuit which applies a correction current to the current detectingterminal, the correction current corresponding to the output voltage atthe secondary side of the transformer, wherein the correction currentgenerated by the correction current generating circuit is applied to acorrection resistor connected to the current detecting terminal for ashift in the voltage applied to the current detecting terminal.
 2. Thepower controlling semiconductor device according to claim 1, furthercomprising: a control terminal which receives the output voltagedetection signal; and an overcurrent limiting voltage generator whichgenerates a predetermined overcurrent limiting voltage, wherein theturning-off signal generating circuit comprises: a voltage selectorwhich selects a lower one of a voltage corresponding to the voltage atthe control terminal and the overcurrent limiting voltage as the upperlimit current detecting voltage; and a voltage comparing circuit whichcompares the upper limit current detecting voltage selected by thevoltage selector with the voltage corresponding to the voltage appliedto the current detecting terminal.
 3. The power controllingsemiconductor device according to claim 1, wherein the transformercomprises a auxiliary winding, and the correction current generatingcircuit comprises: a sample holding circuit which holds a voltagecorresponding to a voltage induced in the auxiliary winding or a dividedvoltage from the induced voltage during a demagnetization period of thetransformer after turning-off of the switching device; and avoltage-current converting circuit which generates a currentcorresponding to the voltage held in the sample holding circuit.
 4. Thepower controlling semiconductor device according to claim 3, wherein thecorrection current generating circuit generates the correction currentin response to the output voltage detection signal, the voltage inducedin the auxiliary winding, or the divided voltage from the inducedvoltage, such that, if a level of the output voltage detection signalincreases from a first level to a second level, the correction currentis adjusted to decrease the level of the output voltage detection signalto the first level or below.
 5. The power controlling semiconductordevice according to claim 1, wherein the turning-on signal generatingcircuit comprises an oscillating circuit which generates an oscillationsignal at a predetermined frequency, and the oscillating circuitgenerates the oscillation signal at a frequency corresponding to thevoltage at the control terminal receiving the output voltage detectionsignal.
 6. A switched-mode power supply comprising: the powercontrolling semiconductor device according to claim 1; a transformer; aswitching device which is connected in series to a primary side windingof the transformer; a current-voltage converting resistor which isconnected in series to the switching device and applies a voltageconverted from a current to the current detecting terminal; an outputvoltage detector which detects an output voltage at a secondary side ofthe transformer and transmits a detection signal to the powercontrolling semiconductor device; and a correction resistor which isconnected between the current detecting terminal and one terminal of thecurrent-voltage converting resistor.
 7. A method of designing aswitched-mode power supply comprising the power controllingsemiconductor device according to claim 3, the method comprising:determining the number of windings and a voltage dividing ratio of afirst resistor to a second resistor such that an input voltage isapplied to a terminal and generates a first correction current at afirst output voltage and a second correction current at a second outputvoltage, an induced voltage from the auxiliary winding or a dividedvoltage which is divided from the induced voltage being applied to theterminal, the first resistor and the second resistor being connected tothe auxiliary winding; determining a first current limit and a secondcurrent limit for the switching device to generate a first outputcurrent at the first output voltage or a second output current at thesecond output voltage in the secondary side of the transformer; anddetermining resistances of a current-voltage converting resistor and acorrection resistor such that the resistances serve as the first andsecond current limits for the switching device if the first and secondoutput voltages and the upper limit current detecting voltage serve asovercurrent limiting voltages, the current-voltage converting resistorand the correction resistor being connected to the current detectingterminal.
 8. A method of designing a switched-mode power supplycomprising the power controlling semiconductor device according to claim5, the method comprising: determining a first upper limit currentdetecting voltage at a first output voltage and a first output currentat the secondary side of the transformer; determining the resistance ofa current-voltage converting resistor connected to the current detectingterminal such that the first upper limit current detecting voltage isequal to the first output voltage at the first output current in thesecondary side of the transformer; determining a second upper limitcurrent detecting voltage equal to a second output voltage at a secondoutput current in the secondary side of the transformer; and determiningthe resistance of the correction resistor such that the second upperlimit current detecting voltage is equal to the second output voltage atthe second output current in response to the correction currentcorresponding to the output voltage applied to the secondary side of thetransformer by a correction current generating circuit and such that thefrequency for turning on or off a switching device is held at or below apredetermined value.
 9. A power controlling semiconductor device forgeneration of a drive pulse to turn on or off a switching device whichapplies an intermittent current to a primary side winding of atransformer in response to a voltage proportional to a present currentflowing through the primary side winding of the transformer and to anoutput voltage detection signal from a secondary side of thetransformer, the semiconductor device comprising: a current detectingterminal to which the voltage proportional to the current through theprimary side winding is applied; a correction resistor which adjusts aresistance of a resistor connected to the current detecting terminal; anovercurrent detecting circuit which compares a voltage corresponding tothe voltage applied to the current detecting terminal with an upperlimit current detecting voltage to detect an overcurrent in thesecondary side of the transformer; a turning-on signal generatingcircuit which generates a turning-on signal for periodically turning onthe switching device; a turning-off signal generating circuit whichgenerates a turning-off signal for turning off the switching device inresponse to the overcurrent detected by the overcurrent detectingcircuit; and a correction current generating circuit which applies acorrection current from the current detecting terminal to the correctionresistor and the resistor connected to the current detecting terminal,the correction current corresponding to the output voltage at thesecondary side of the transformer, wherein, in order to detect theovercurrent, the overcurrent detecting circuit compares the upper limitcurrent detecting voltage with a voltage corresponding to an appliedvoltage which is applied to the current detecting terminal and which isshifted by a correction current generated by the correction currentgenerating circuit through the correction resistor.
 10. A switched-modepower supply comprising: the power controlling semiconductor deviceaccording to claim 9; a transformer; a switching device which isconnected in series to a primary side winding of the transformer; acurrent-voltage converting resistor which is connected in series to theswitching device and applies a voltage converted from a current to thecurrent detecting terminal; and an output voltage detector which detectsan output voltage at a secondary side of the transformer and transmits adetection signal to the power controlling semiconductor device, whereinthe switching device is turned on or off by a drive pulse generated bythe power controlling semiconductor device.